REFLECTIVE ANTENNA WITH MULTI-FREQUENCY PHASE CORRECTION FOR SPACE APPLICATIONS AND DESIGN METHOD OF

专利摘要:
This invention consists of a reflector antenna operating in several frequency bands with improved radiation patterns in all bands. The antenna has the reflective surface covered with one or more layers of dielectric with small conductive elements. A small part of the conductive elements printed on the surface of the reflector, selected by an algorithm that depends on several geometric parameters, are designed to invert the phase of the reflected field in the higher frequency bands, without modifying the characteristics of the antenna in the lower frequency bands. The result is a frequency selective phase correction that allows the beam radiated or received by the antenna to be shaped independently in various frequency bands. This invention makes it possible to design antennas to generate multi-cellular coverage from a satellite, ensuring an identical size of the coverage areas for the upstream and downstream beams. (Machine-translation by Google Translate, not legally binding)
公开号:ES2791798A1
申请号:ES202030819
申请日:2020-07-31
公开日:2020-11-05
发明作者:Garcinuno Jose Antonio Encinar；De Rioja Del Nido Jose Daniel Martinez；De Rioja Del Nido Eduardo Maria Martinez；Diaz Rafael Florencio；Boix Rafael Rodriguez
申请人:Universidad Politecnica de Madrid；
IPC主号:

专利说明:

[0002] Reflector antenna with multi-frequency phase correction for space applications and design method of the same
[0004] TECHNICAL SECTOR
[0005] This invention is part of the radiocommunication technology and space technology sectors.
[0007] BACKGROUND OF THE INVENTION
[0008] Science mission and communication satellites use high gain reflective antennas to transmit and receive microwave frequency, or millimeter wave signals, to and from ground communication stations or user terminals. In the antennas used in satellites, both for communications and for scientific missions, it is usual to use the same reflector for two or more frequency bands. In scientific missions, a large reflector is used to transmit and receive data to / from the ground in the bands called: S (2 GHz), X (7-8 GHz) and Ka (32-34 GHz), as can be seen in [R. Mizzoni, "The Cassini High Gain Antenna (HGA): a survey on electrical requirements, design and performance," IEE / SEE Seminar on Spacecraft Antennas, London, UK, 1994, pp. 6 / 1-610] and in [N. Chahat, J. Sauder, M. Mitchell, N. Beidleman and G. Freebury, "One-Meter Deployable Mesh Reflector for Deep-Space NetWork Telecommunication at X-Band and Ka-Band," IEEE Trans. on Antennas and Propagation, vol. 68, no. 2, pp. 727-735, Feb.2020].
[0010] Geostationary satellites that are used to provide cellular coverage for broadband Internet access with frequency and polarization reuse, considering the configuration of a feeder for each beam, require four reflector antennas of more than two meters in diameter to generate complete coverage, made up of a high number of beams (between 50 and 100); where each of the reflectors is fed by a group of horn antennas, in which each horn transmits and receives the signals associated with each beam, as can be seen in several documents of the prior art, such as [US2007018900A1 Patent "Multi-beam and multi-band antenna system for communication satellites"] and [SK Rao, "Advanced antenna technologies for satellite communications payloads", IEEE Trans. Antennas Propag., Vol. 63, no. 4, pp. 1205-1217, April 2015]. Each reflector operates on the transmission (17.5-20.2 GHz) and reception (29-30 GHz) frequencies to generate a quarter of the beams in a given polarization and frequency band. The problem that appears is that the receiving beams on the satellite (upward beam) are narrower and have higher gain (about 3 dB) than the downward beams (transmission from the satellite), because the frequency is higher and the reflective surface is the same. This effect creates a problem in coverage areas as it would require ground user antennas to emit much higher power if they are near the edge of coverage (EOC), than in the center of the area (the difference in levels would be more than 7 dB, when the recommended value is around 4 dB). To solve this problem, several solutions have been proposed, which are discussed below.
[0012] A first solution to reduce the gain is to shape the surface of the reflector, generally at the edges. A version of this conformation consists of making several steps near the edge of the reflector ["Stepped-reflector antenna for satellite communication payloads" US7737903B1 Patent, June, 2005]. This technique has several drawbacks. The first is that the improvement in performance in one frequency band is achieved at the cost of a small deterioration in the other band. Secondly, the shaping of reflectors for space requires the manufacture of a specific mold for each mission, with great precision in manufacturing and thermal stability (typically it is done in invar to avoid dilations in the manufacturing processes of the antenna in fiberglass). carbon).
[0014] A second solution consists of a more complex design of the antenna system, including an optimization of the feeder profile [US2003142014A1 Patent “Dual-band multiple beam antenna system for communication satellites ”], which can be combined with a dual-reflector system [“ Dual-band wave beam equalization side-fed offset cassegrain antenna and realization method thereof, Chinese patent CN102800993B, Nov.
[0015] 2014], In these cases, the horns are designed so that their beamwidth and phase center are frequency dependent. The center horn is positioned so that the phase center is at the focus of the paraboloid at the transmitting frequency, the phase center being shifted at the receiving frequency. So the beam is slightly out of focus at the receiving frequency (upward beam). This reduces the gain at higher frequencies, although it is difficult to keep the required beamwidth at -4 dB of the maximum, which is the so-called EOC contour. In addition, it can increase the level of secondary lobes, which would generate more interference with other beams in the coverage, which can be more significant as the number of beams is greater. On the other hand, a more complex design of the horns entails a greater length of the same, increasing the weight and volume of the power supply network.
[0017] Third, in the patent [T. K. Wu, "High performance multi-band frequency selective reflector with equal beam coverage", Patent US2003164803A1, Sept. 2003] it was proposed to add an edge to the reflecting antenna with a frequency selective surface (FSS in the English literature), which absorbs the signal received at the satellite for the higher frequencies (uplink), while not affecting the performance of the antenna in transmission at the lowest frequencies. This technique has the drawback of being more difficult to manufacture, as different materials have to be used on the edges of the reflector. This technique of correcting or equalizing the beamwidth with the frequency allows only to reduce the effective size of the antenna for higher frequencies, because the signal incident on the outer ring is dissipated in an absorbent material, but it lacks flexibility to shape the bundle, or reduce the secondary lobes.
[0018] A similar technique is that proposed in the patent [US2004017332A1 "Multiple beam antenna using reflective and partially reflective surfaces"], which consists of in a reflector formed by a conductive central area and several rings that surround the central area, made of materials that partially reflect the signal at higher frequencies. This reduces the effective size of the antenna for higher frequencies. The result is a reduction in gain, but at the cost of this increases the level of secondary lobes in the higher frequencies.
[0020] To generate multi-beam cellular coverage from the satellite, “reflectarray” antennas have also recently been proposed, which would allow the total number of antennas on the satellite to be reduced from four to two antennas [M. Zhou, S. B. Serensen, Y. Brand and G. Toso, "Doubly Curved Reflectarray for Dual-Band Multiple Spot Beam Communication Satellites," IEEE Transactions on Antennas and Propagation, vol. 68, no. 3, pp. 2087-2096, March 2020], since each “reflectarray” antenna is capable of generating two beams in circular orthogonal polarization for each feed horn. In this way, each "reflectarray" antenna is capable of generating twice as many beams as a conventional reflector and, therefore, can generate half of the cellular coverage required for a given geographic region. However, in the results presented in a 60-cm diameter model, it is observed that the beam is narrower in reception (with a gain between 1 and 2 dB greater than in transmission), since no correction has been made. of the diagrams on the reception frequency. This difference can increase dramatically for an antenna of real dimensions (around two meters in diameter), since the required gain is much higher.
[0022] A "reflectarray" antenna consists of a grouping of phase shifter cells distributed in a regular grid. In the "reflectarray" a phase adjustment of the reflected field is introduced into all cells of the "reflectarray" to produce a collimated or shaped electromagnetic beam when illuminated by a primary feeder. The phase shifting cells used in the so-called “printed” “reflectarrays” are formed by conductive elements printed on one or more layers of dielectric substrate located on a conductive plane, so that the phase shift produced by each cell is adjusted by varying dimensions [DM Pozar and TA Metzler, “Analysis of a reflectarray antenna using microstrip patches of variable size”, Electr. Lett. Vol. 29, No. 8, pp. 657-658, April 1993], or the angle of rotation of the printed conductive elements [J. Huang, "A Ka-Band Microstrip Reflectarray with Elements Having Variable Rotation Angles", IEEE Trans. Antennas Propagat., Vol. 46, No. 5, pp. 650-656, May 1998]. In a "reflectarray", the phase shifter cells are designed to collimate or shape the radiation pattern in a frequency band, and generally in other frequency bands the behavior of the antenna is totally distorted.
[0024] In addition to “reflectarrays” on flat surfaces, in the patent [M. E. Cooley, T. J. Chwalek, "Method for improving pattern bandwidth of shaped beam reflectarrays", US006031506 Patent, 2000] a "reflectarray" was proposed on a parabolic surface to design shaped beam antennas with improved bandwidth. However, this type of antenna operates only in one frequency band.
[0026] In the prior state of the art, a large number of works can be found on "reflectarray" antennas designed to operate on two or more frequencies. A design technique for a single-layer multi-band “reflectarray” was described in [M. Borgese, F. Costa, S. Genovesi and A. Monorchio, "An iterative design approach for multi-band single-layer reflectarrays," 2015 9th European Conference on Antennas and Propagation (EuCAP), Lisbon, 2015, pp. 1-6]. In this work, the design was applied to three frequencies (3.9 GHz, 7.5 GHz and 12.5 GHz), it used phase shifting cells formed by three concentric rings and only simulation results were presented, without experimental validation. The main limitation of this technique is that phase shifter cells, or "reflectarray" cells, only operate in linear polarization.
[0028] In the work published in [J. Shaker and M. Cuhaci, “Multi-band, multipolarization reflector-reflectarray antenna with simplified feed system and mutually independent radiation patterns,” IEE Proc. - Microw. Antennas Propag., Vol. 152, no. 2, pp. 97-102, 2005], an antenna system combining a “reflectarray”, a frequency selective surface and a conventional reflector for multi-frequency operation in dual-polarization applications at two frequencies was proposed and demonstrated. The proposed configuration is capable of achieving independent radiation patterns for each frequency, at the cost of considerably increasing the volume and mass of the antenna and having to use different feeders for each frequency band, which means a serious limitation for on-board antennas. on satellites.
[0030] Another “reflectarray” antenna configuration that operates in two frequency bands has been implemented by placing a Ka-band “reflectarray” on top of an X-band “reflectarray” [Chaharmir, M.R .; Shaker, J .; Legay, H., "Dual-band Ka / X reflectarray with broadband loop elements," in Microwaves, Antennas & Propagation, IET, vol. 4, no. 2, pp. 225-231, Feb.
[0031] 2010]. Subsequently, a multi-band “reflectarray” has been proposed that transmits and receives in the Ka and X bands [Chaharmir, M.R .; Shaker, J., "Design of a Multilayer X- / Ka-Band Frequency-Selective Surface-Backed Reflectarray for Satellite Applications," IEEE Transactions on Antennas and Propagation, vol. 63, no. 4, pp. 1255-1262, April 2015], which uses a different “reflectarray” for each band and a frequency selective surface (FSS) that allows the two frequency bands to be separated. In both cases, the resulting antenna is made up of many layers and is very thick, not being suitable for space antennas, due to the increased volume and weight on the satellite.
[0033] Another type of plane “reflectarray” that operates in two different frequency bands (12 GHz and 20 GHz) with independent beams in each polarization was described in [E. Martinez-de-Rioja, JA Encinar, M. Barba, R. Florencio, RR Boix and V. Losada, "Dual Polarized Reflectarray Transmit Antenna for Operation in Ku- and Ka-Bands With Independent Feeds," IEEE Transactions on Antennas and Propagation , vol. 65, no. 6, pp. 3241-3246, June 2017]. In this work, the "reflectarray" cells were formed by independent sets of parallel dipoles for each polarization displaced between yes half period, in order to provide independent phase control on each linear polarization. The multi-frequency reflectarray cell comprises two sets of three (or five) coplanar parallel dipoles printed on a first ground plane dielectric layer, and three additional parallel dipoles on a second layer of dielectric material arranged on top of the first. . The dipoles in the first dielectric layer are used to adjust the required phase at the lowest frequency (12 GHz), while those in the second layer are adjusted to provide the required phase at the highest frequency (20 GHz). The same type of “reflectarray” cell has also been applied to design a “reflectarray” that operates at 20 GHz and 30 GHz [E. Martinez-de-Rioja, JA Encinar, R. Florencio and RR Boix, “Reflectarray in K and Ka Bands with Independent Beams in Each Polarization”, IEEE APS 2016 Fajardo, Puerto Rico], although no results have been presented in this work experimental. In both works, flat multilayer “reflectarray” antennas are studied, capable of operating in two frequency bands, where the sets of dipoles printed in each cell of the “reflectarray” have been optimized to shape the beam at each frequency, with which the levels The gains obtained are different at each frequency.
[0034] It should be noted that a "reflectarray" antenna has in common with this invention the fact that conductive elements are employed to perform a phase adjustment in the reflected field. However, the concept is different, since the phase adjustment in a "reflectarray" is performed in all cells of a regular grating to shape, focus or deflect the beam. The dimensions and, where appropriate, angle of rotation, of the conductive elements printed in each and every one of the cells of the “reflectarray” must be optimized to achieve the required phase values with a small margin of error at all frequencies of antenna operation. This is an intensive computational process that requires a lot of computation time, due to the high number of cells that make up a reflectarray antenna (generally more than 20,000 for a communications satellite antenna).
[0035] The "reflectarrays" described in the prior art can be used in multi-beam antennas, but like reflectors, they have the drawback of presenting different beam widths in the transmission and reception frequency bands when these are sufficiently separate, as is the case with satellite antennas operating in the Ka band (20 GHz and 30 GHz). This is the problem that the invention solves. In fact, the present invention can be used in both reflector type and "reflectarray" type antennas. In the case of “reflectarray” antennas, only a small percentage of the “reflectarray” cells (around 10%) will be replaced by others that perform a certain phase change, which greatly simplifies the implementation of this invention, since that only requires redesigning a small number of cells, and does not entail any increase in manufacturing processes and costs.
[0037] DESCRIPTION OF THE INVENTION
[0038] The invention relates to a multi-frequency phase-corrected reflective antenna for transmitting and receiving signals in various frequency bands, exhibiting improved radiation characteristics at higher frequencies, according to claim 1, and with a method for obtaining said antenna according to claim 11. Preferred embodiments of the antenna and the method are defined in the dependent claims.
[0040] The reflector antenna with multi-frequency phase correction that transmits and receives signals in at least two frequency bands to improve the radiation patterns in the bands, comprises a reflector and a primary feeder configured to illuminate the reflector, the reflector being formed by a structural base covered by one or more layers of conductive and dielectric materials, including conductive elements on the surface of at least one layer of dielectric material. The frequency bands in which the antenna operates are named by their ordinal, so that the first band corresponds to the lower frequency band, the second band to the immediately higher frequency band, and so on. As an example, we can consider an antenna that operates in three frequency bands, the central frequencies of each band being: 12 GHz, 20 GHz and 30 GHz. The first band would be the lower band (centered on 12 GHz), the second band would be the immediately higher one ( centered at 20 GHz), and the third band (or upper band) would be centered at 30 GHz. The main aspect of the present invention is that at least a part of the conductive elements printed on the surface of at least one layer of material dielectric are configured to invert the phase of the field reflected by said conductive elements in at least one of the higher order frequency bands (higher frequencies) in which the antenna operates, without producing an effect on the phase of the field reflected in the first band, or lower frequency band. To produce frequency selective phase correction, that is, correcting the phase in one or more higher order frequency bands without affecting the phases in the first frequency band, each of the phase inverting conductive elements has to have dimensions equal to or less than half a wavelength in "said at least one higher order frequency band" in which the phase is inverted, and at the same time, equal to or less than 0.3 wavelengths in the band of immediately lower frequency. Furthermore, the position and distribution of said phase inverting conductive elements (which change the phase 180o in the at least one higher-order frequency band) are defined by a random distribution with a variable density that adjusts to form a radiation pattern. in at least one higher-order frequency band (higher frequencies) without modifying the radiation pattern in the first frequency band (lower frequencies), to improve radiation patterns in all frequency bands.
[0042] In a first preferred embodiment, the antenna consists of a conventional reflector covered with one or more layers of dielectric material, where at least one of the layers contains metallizations in certain previously calculated positions that produce a phase change in at least one band of higher frequency, without modifying the phases in the lower order bands (lower frequencies), as described above.
[0043] In a second preferred embodiment, the antenna contains conductive elements distributed in cells in a regular lattice on the surface of at least one layer of dielectric material, so that said conductive elements comprise a shape, size and angle of rotation with respect to the lattice previously calculated to generate an initial phase distribution in the reflected field necessary to collimate or shape the radiated beam in all transmit and receive bands of the antenna. In this preferred embodiment, a part of the cells of the lattice comprises phase inverting conductive elements, configured to carry out a 180 ° phase change in the higher order frequency bands, with respect to the initial phases generated to collimate or shape the beam.
[0045] A third preferred embodiment consists of a "reflectarray" type reflector antenna for circular polarization with multi-frequency phase correction that includes at least a first and a second group of conductive elements in each cell of a regular grating on the surface of at least one layer of dielectric material. So a rotation angle of the first group of conductive elements is used to adjust the phase in the first frequency band (or lower frequency band) and a rotation angle of the second group of conductive elements is used to adjust the phase in the second. frequency band (higher frequency), using the technique known as sequential rotation independently for each group of conductive elements, taking into account that each group of conductive elements in the cells of the reflectarray is associated with a frequency band. In this embodiment, the conductive elements of each group have a shape and size previously calculated to introduce a 180 ° phase shift between the two components of the electric field reflected by the conductive element in each of the operating frequency bands of the antenna. Once the previous condition of 180 ° phase shift between the two reflected field components, which is a premise in the sequential rotation technique, is fulfilled, the reflected field of circular polarization in each frequency band will have a phase shift in each cell of the antenna What is it proportional to twice the angle of rotation of each group of conductive elements.
[0047] The conductive elements of each group are rotated in the lattice at angles equal to half the phase required in each frequency band of the antenna to collimate or shape the beam in circular polarization, when a feeder operating in polarization is used. circulate in various frequency bands. In this preferred embodiment, a part of the conductive elements of the second group, which are associated with the second frequency band (with higher frequencies than in the first band) have different dimensions that have been previously calculated to invert the phase of the field. reflected in circular polarization in the second frequency band, with respect to the phase initially calculated to collimate or shape the beam.
[0049] As a particular case of the third preferred embodiment of a reflector antenna for circular polarization with multi-frequency phase correction, two symmetrical arcs with a rotation angle aArc are used as the conducting element of the first group, used to control the phase in the first group. band (or lower frequency band), and two orthogonal sets of three parallel dipoles, each of them printed on a dielectric layer, with a rotation angle aD¡P, as the conducting element of the second group, which is used to adjust the phase in the second frequency band, or higher frequency.
[0050] In any of the previously described embodiments, it is possible to maintain a central area of the reflector without phase correction, and have the conductive elements that invert the phase in the frequency bands in which the antenna operates, excluding the first band, distributed only in outer crowns, where each outer crown and each configuration of phase inverting conductive elements (which invert the phase in the upper frequency bands) are defined for each frequency band in which the correction is made.
[0051] In any of the previously described embodiments, a phase correction technique consists of having the conductive elements that invert the phase in the frequency bands in which the antenna operates, excluding the first band, distributed in random positions within one or more outer crowns, where the random distribution of the phase corrected elements in each crown is defined by an algorithm that includes at least two adjustment parameters. The algorithm to define the positions of the elements with phase correction is applied independently for each frequency band in which the correction is made.
[0053] In any of the preferred embodiments described, the conductive elements that invert the phase in the frequency bands in which the antenna operates, excluding the first band, may be randomly distributed with increasing density from the center of the antenna to the edge.
[0054] In any of the preferred embodiments described, the antenna can have a flat, parabolic, spherical, or cylindrical reflective surface.
[0056] In any of the preferred embodiments described, the reflector antenna can be constructed of materials qualified for space applications, such as pre-impregnated carbon fiber with resins, copper-coated Kapton, copper-coated Kapton-Germanium, pre-impregnated quartz fibers. with low loss resins and qualified adhesives for space applications, as well as a combination thereof.
[0058] In any of the preferred embodiments described, the antenna can have a parabolic reflective surface, and be designed to generate multi-cellular coverage from a geostationary satellite with a phase correction in the second frequency band, which is used for reception, calculated to obtain the same beamwidth and the same gain in the first frequency band, which is used for transmission, and in the second frequency band, used for reception.
[0059] According to another aspect of the invention, a method is presented to obtain a reflecting antenna that transmits and receives signals in at least two frequency bands, which are called by their ordinal, so that the first band corresponds to that of lower frequencies , the second band at the immediately higher frequencies, and so on. The antenna includes conductive elements distributed in cells on the surface of at least one layer of dielectric material. The method comprises a first step, defining the specifications of the radiation patterns in the antenna operating frequency bands, ordered from the lower frequency (first band) to the higher frequency band (fn), and a second step, of antenna design without phase correction that generates the radiation patterns in the defined directions according to the preset specifications in all the operating frequency bands of the antenna. Once the two preliminary steps of the conventional design of the antenna have been carried out, the method allows a multi-frequency phase correction to be carried out to improve the radiation patterns in all bands.As aspects of the invention, the method comprises the following steps:
[0060] a) Define a regular grid on the reflecting surface of the antenna and define the first frequency band as the lower band (//, / = 1). b) Obtain some positions and density of some cells in the regular grid in which the phase will be inverted in the immediately superior frequency band (// + 1), according to a selection of random distribution, which is a function of the distance of the cell to the edge of the reflector and that is adjusted with at least two parameters.
[0061] c) Introduce in the cells selected in the previous point a 180o phase change, or phase inversion, for the reflected field in the immediately higher frequency band, without changing the phase in any of the lower frequency working bands, and calculating the radiation patterns in the two frequency bands called lower (//) and immediately upper (// + 1), after introducing the phase inversion in the selected cells. d) Compare the radiation patterns obtained after the phase correction performed in step c) with the diagrams obtained in the second step before the phase correction, and with the predefined specifications in the first step for the radiation patterns at each frequency.
[0062] In the event that the diagram specifications are not met at both frequencies, you must:
[0063] e) Modify the parameters used to adjust the distribution of cells in which the phase inversion is performed, obtain new cell positions with phase inversion at the immediately higher frequency and repeat steps c) and d).
[0064] f) Define the geometry and adjust the dimensions of the conductive elements in each cell selected above to invert the phase in the immediately higher frequency band (Z + 1), without altering the phases in the lower frequency band (Z), to an angle of incidence that corresponds to the center of the reflecting surface. In the case where the antenna must operate in an additional higher frequency band, the previous immediately higher band is redefined as the lower band, and the new band is considered as the immediately higher frequency. Then the following steps are performed:
[0065] g) Obtain new positions and density of cells in the lattice according to a random distribution with a density that depends on the distance from the cell to the edge of the reflector and that is adjusted with at least two parameters; to introduce in said cells a phase inversion (or phase shift of 180 °) in the reflected field for the new immediately higher frequency band, without affecting the phase distribution in the lower bands.
[0066] h) Calculate the radiation patterns in all the considered frequency bands after introducing the phase inversions in the selected cells, compare them with the patterns before correction and with the predefined specifications for the radiation patterns in each frequency band. frequency.
[0067] If the diagram specifications are not met in all considered frequency bands:
[0068] i) Modify the parameters used to adjust the distribution of cells in which the phase inversion is carried out for the immediately higher frequency band and repeat steps g) and h). j) Define the geometry and adjust the dimensions of the phase inverting conductive elements in the immediately higher frequency band, without altering the phases in the lower frequency bands, for the angle of incidence that corresponds to the center of the antenna.
[0069] Once the previous steps have been carried out for all the frequency bands, we proceed to:
[0070] k) Arrange the elements previously defined in the positions obtained to carry out the phase inversion in each frequency band, and calculate the radiation patterns in all the frequency bands.
[0071] l) Optimize the dimensions of all the conductive elements used to carry out the phase inversion, through an optimization routine that iteratively calls an electromagnetic modeling routine of the conductive elements in a periodic environment, until the specifications are met in all bands frequency. m) From the dimensions of all the conductive elements obtained in the optimization process, the masks are generated for the manufacture by photoengraving of the dielectric layers with the optimized conductive elements.
[0072] In the second preferred embodiment, the method also comprises as a step prior to step b): adjusting the dimensions of the conductive elements distributed in a regular lattice on the surface of at least one layer of dielectric material, to generate the phase distribution in the reflected field necessary to collimate or shape the radiated beam in all transmit and receive bands of the antenna. Subsequently, in steps from j) to), only the dimensions of a part of the conductive elements of the grating are adjusted to introduce the phase inversion in the higher order frequency bands, with respect to the phase initially calculated to collimate. or shape the bundle.
[0073] In the third preferred embodiment for a circular polarization antenna, the method also comprises as a step prior to step b):
[0074] - Define the geometry and adjust the dimensions of at least a first group and a second group of conductive elements in each cell of a uniform lattice on the surface of at least one layer of dielectric material, to introduce a phase change of 180 ° between the two components of the electric field reflected by the conductive element in each of the antenna's operating bands.
[0075] - Adjust the angle of rotation of the first and second group of conductive elements using the technique known as sequential rotation, to obtain the angles of rotation of the elements of the first group to conform the beam to the lower frequency band, and the angles of rotation of the elements of the second group and of the following groups, if any, to shape the beam in the second frequency band and in the following bands, respectively, when the feeder operates in circular polarization at various frequencies.
[0076] - Adjust, in steps from j) to), the dimensions of the conducting elements of the second group and subsequent groups, if any, used to control the phase in the second frequency band and subsequent ones, if any, to introduce a phase inversion in the reflected field in circular polarization in the second frequency band and subsequent ones, if any, with respect to the phase initially calculated in the second step to collimate or shape the beam in each frequency band.
[0078] In any of the preferred embodiments described, the cells selected in the method for introducing a 180 ° phase shift (phase inversion) for the higher frequency bands can only be found in an outer corona, which is specifically defined for each frequency band.
[0079] BRIEF DESCRIPTION OF THE DRAWINGS
[0080] A series of drawings that help a better understanding of the invention and that are expressly related to various embodiments of said invention, presented as a non-limiting example, are briefly described below:
[0082] Figure 1. Perspective of a reflector antenna (usually parabolic) operating in two frequency bands with phase correction in the upper band, according to a first embodiment of the invention.
[0084] Figure 2. Perspective of a planar "reflectarray" antenna operating in two frequency bands with phase correction in the upper band, according to a second embodiment of the invention.
[0086] Figure 3. Perspective of a parabolic reflectarray antenna operating in two frequency bands in circular polarization with phase correction in the upper band according to a third embodiment of the invention.
[0088] Figure 4. Representation of the reflector layers for a group of 3x3 cells that includes two types of square conducting elements of different sizes, the largest being those used to introduce a phase correction in the upper frequency band.
[0090] Figure 5. Phase curves of the reflection coefficient for the two elements of different size shown in figure 4.
[0092] Figure 6. Radiation patterns in the XZ plane corresponding to a conventional reflector antenna made of carbon fiber with an aperture diameter of 1.8 m at the frequencies of 19.7 GHz and 29.5 GHz, according to the previous state of the technique.
[0094] Figure 7A. Distribution of phases introduced in the reflector at the frequency of 29.5 GHz when a phase correction is introduced in an outer corona according to a random distribution.
[0095] Figure 7B. Radiation patterns in the XZ plane for a 1.8 meter reflector with the phase correction shown in Figure 7A at the frequencies 19.7 GHz and 29.5 GHz.
[0097] Figures 8A-8B. Distribution of phases introduced into the reflector at the frequency of 29.5 GHz (Fig. 8A) and radiation patterns in the XZ plane (Fig. 8B) at the frequencies of 19.7 GHz and 29.5 GHz, when switched the phase in a corona with a random distribution of increasing density towards the edge of the reflector.
[0098] Figure 9A-9B. Distribution of phases introduced into the reflector at the frequency of 29.5 GHz (Fig. 9A) and the radiation patterns in the XZ plane (Fig. 9B), when the phase is changed according to a random distribution with increasing density towards the reflector edge.
[0100] Figure 10. Perspective view of a group of "reflectarray" cells used to introduce a phase correction in the upper frequency band in a "reflectarray" operating in two frequency bands.
[0102] Figures 11A and 11B. Phase values of the reflection coefficient at the frequencies of 19.7 GHz and 29.5 GHz, respectively, as a function of the lengths of the dipoles in each layer for the “reflectarray” cells shown in figure 10.
[0104] Figures 12A-12C. Phase distribution in a "reflectarray" antenna at the frequencies of 19.5 GHz (Fig. 12A) and 29.5 GHz (Fig. 12B), and the corresponding radiation patterns (Fig. 12C) in the XZ plane, for a 1.8 m diameter "reflectarray" without phase correction, according to the prior art.
[0106] Figures 13A-13B. Phase distribution at 29.5 GHz (Fig. 13A) after correcting for phase distribution in the "reflectarray" shown in Fig. 12B, and radiation patterns in the XZ plane (Fig. 13B) at 19.7 GHz and 29.5 GHz, according to a second embodiment of the invention.
[0107] Figures 14A-14B. Top and perspective views of a circular polarized “reflectarray” cell to adjust the phase according to the sequential rotation technique in two frequency bands.
[0109] Figure 14C. Top view of a circularly polarized “reflectarray” cell that reverses the phase in the upper frequency band.
[0111] Figures 15A-15B. Phase values of the X and Y components, respectively, of the reflection coefficient at 29.5 GHz, when the lengths of the dipoles in each layer are independently varied in the "reflectarray" cells shown in Figure 14A.
[0113] Figures 16A-16B. Phase distribution in circular polarization at the frequency of 29.5 GHz (Fig. 16A) and radiation patterns in the XZ plane (Fig. 16B) at 19.7 GHz and 29.5 GHz, for a parabolic reflectarray ( Fig. 3) according to a third embodiment of the invention.
[0115] Figure 17. Flow diagram of the design technique for multi-frequency phase correction.
[0117] DESCRIPTION OF A PREFERRED EMBODIMENT
[0118] The following description of the preferred embodiments are merely examples and are not intended in any way to limit the invention, its applications or uses.
[0120] This patent proposes several embodiments of reflective antennas used to transmit and receive signals in various frequency bands, including a reflector (10) formed by a multilayer structure (11) to provide multi-frequency correction in radiation patterns already at the same time the mechanical rigidity to avoid distortions on the surface, a feeder (12) that emits and receives electromagnetic signals in different frequency bands in the direction of the center of the reflector (10), and that is assembled to the structure (11) by means of an arm (13). The surface of the reflector (10) is the external face of the structure (11) formed by one or more layers of dielectric material with conductive elements (14, 15; 21, 22, 23, 24; 33, 34A-B, 35) of dimensions smaller than half a wavelength at the highest operating frequency, which are glued to other structural layers. The conductive elements (14, 15; 21, 22, 23, 24; 33, 34A-B, 35) are regularly distributed in some cells (18, 19; 25, 26; 31, 32) that result from defining a grid regular (16) on the surface of the reflector. This regular grid has only been represented in area (16), but it represents the distribution of the cells over the entire surface of the reflector, in which not all the references of the elements it contains have been included to avoid agglomeration of references and thus facilitate understanding of the invention. The division of the reflector surface into quasi-periodic cells is carried out to be able to analyze and design the reflector by characterizing the resulting cells. Each cell comprises one or more conductive elements printed on one or more layers of dielectric substrate on a conductive plane. When the electromagnetic field coming from the feeder affects one of these cells, the locally reflected field in said cell is calculated in a conventional way, assuming that it is in a periodic environment, to take into account the coupling of electromagnetic fields with the conductive elements of the adjoining cells. The phase shift produced in the field reflected by each cell is controlled by the dimensions of the conductive elements contained in the cell.
[0122] A small part of the cells made up of conductive elements are designed to invert the phase (i.e. introduce a 180o phase shift) of the reflected field on the reflector surface in a higher frequency band, without altering the phase of the reflected field in the lower bands. In Figure 1, the conductive elements that invert the phase are large patches (15), while the rest of the conductive elements are small patches (14) in Figure 1, which do not produce any appreciable change in the phase with respect to an antenna designed according to the prior art. The conductive elements that invert the phase in the upper frequency band are located in some cells, as will be described, distributed randomly in a regular grid (16) on the surface of the reflector (10), in order to adjust the width of do and the gain, so that they coincide with these same values in the lower frequency band. The reversal of the phase of the reflected field in a few randomly distributed cells produces a correction of the overall phase distribution on the surface of the reflector, which can be modulated, by controlling the density and position of the cells with phase inverting conductive elements. It should be noted that phase inverting conductive elements are frequency selective, so that they only invert the phase in the defined frequency band, while they do not produce changes in the phase of the reflected field for lower frequency bands. The effect of inverting the phase only in one frequency band is achieved thanks to the fact that the phase inverting conductive elements have dimensions that are less than a third of the wavelength in the lower frequency bands, so that they do not affect these bands. The proposed phase correction can be applied both to conventional reflectors (Fig. 1) and to “reflectarray” type antennas (Fig. 2), in which a phase distribution is introduced through a plurality of phase shifting cells (25), which only They are phase shifting and non-inverting, containing printed conductive elements (21, 22) regularly distributed in a grid (16) on the surface of the reflector, to focus or shape the beam, said "reflectarrays" being able to be manufactured on a flat surface (Fig. 2) or concave, for example parabolic (Fig. 3) or spherical. Each of the cells in the "reflectarray" produces a controlled phase shift in the electromagnetic field reflected in the cell, and therefore, we call them phase shifters. Therefore, the cells of a "reflectarray" imply the offset characteristic. In figure 2, an elliptical broken line (20) has been represented, in which a plurality of phase shifting cells (25) are included, but as in the case of the grid (16), said reference (20) represents the distribution of the phase shifting cells (25) over the entire surface of the reflector. The conducting elements of the phase shifting cells (25) are formed by two groups of printed dipoles (21, 22), some (21) in the direction of the X axis and others (22) in the direction of the Y axis, as is known in the state of the art.
[0123] Figure 3 shows an elliptical broken line (30), similar to (20), but which, in this case, includes a plurality of phase shifting cells (31), similar to cells (25) but with different configuration , as described below. The plurality of phase shifter cells (31) are formed by printed elements (33, 34A-B), as discussed below. The phase correction by means of phase inverting conductive elements can be carried out only in an outer crown (17) or in the entire reflective surface, with a uniform density of phase inverting conducting elements, or with a variable density of said inverting elements depending on the distance to the edges of the reflector, as described below. These capabilities provide greater flexibility for phasing at higher frequencies, which can be used to produce a different beam shaping at each frequency, without the need to shape the reflective surface. That is, the same antenna could be reused for different missions, such as shaped beams on one frequency and multiple beams on another. Another advantage, with respect to the different configurations proposed in the prior state of the art, is that their manufacture is simple, materials and processes verified in space antenna technology are used, and the weight and volume of the antenna with phase correction it is similar to that of an antenna without such correction. The implementation of the phase correction technique is carried out through a systematic design process that requires a low computational cost.
[0124] An application of interest is found in multi-beam antennas used in geostationary satellites to provide cellular coverage for broadband Internet access with frequency and polarization reuse. Currently, four reflector antennas of more than two meters in diameter are used to generate complete coverage, made up of a high number of beams (between 50 and 100) that, projected on the surface of the earth, produce a cellular coverage. Each beam from each of the four antennas is associated with a ground coverage area. For antennas in the so-called Ka band, each reflector operates in a lower frequency band to transmit to the ground (17.5-20.2 GHz) and in a higher one for reception (29-30 GHz), generating in both bands a quarter of the beams of the total coverage. The beams from the same reflector operate in a certain polarization and frequency sub-band, taking into account that the total frequency assigned to users is divided into two frequency sub-bands that alternate from one cell to another to minimize the interference between cells. By transmitting and receiving in different frequency bands using the same reflective surface, the receiving beams (upward beam) are narrower and have higher gain (around 3 dB more) than the downward beams (transmitted from the satellite). The present invention makes it possible to carry out a phase adjustment only in the upper frequency band (reception in the satellite), avoiding complex designs of the feeder or those shaped on the surface of the reflector, so that the beams in reception have the same performance as in transmission. That is, it is achieved that the ground coverage areas have the same size in transmission and reception.
[0126] In order to adjust the beamwidth and gain in the upper frequency band, phase inverting conductive elements (15; 23, 24; 35) are introduced, with a size of the order of half a wavelength in the frequency band. higher frequencies that is selected to produce a phase reversal of the reflected field (or 180o shift) in the higher frequency band, relative to what the reflected field would have on the surface of the conventionally designed reflector. The size of the conductive elements is chosen small enough (less than one third of the wavelength in the lower frequency bands) so that they produce a negligible phase shift in the lower frequency bands. The conductive elements (15) that invert the phase in the upper band are placed in cells (19) randomly distributed on a regular lattice (16) in a crown (17) that begins at the edge of the reflector. The inversion or phase change of 180o, which means a sign change in the signal, in certain cells distributed randomly, produces random variations of sign in the reflected field, which would cancel the radiated field produced by other cells, by a destructive interference process. This correction has the The same effect as an effective reduction in the size of the antenna, only for the highest frequencies, thereby reducing the antenna gain and increasing the beamwidth, until they are equal to the beamwidth and gain obtained at lower frequencies. However, this correction offers much greater flexibility in beam shaping than a simple effective size reduction, since although the distribution of the phase reversed cells is random, more parameters can be added to control the beam shaping to the higher frequencies. These parameters include: 1) variation in the density of points with phase inversion on the reflector surface, 2) width of the corona (17) where the correction is made, 3) Cells with phase inversion can be included in the entire surface with increasing density from the center towards the edges.
[0127] The design procedure, according to the particular embodiment of the invention, to implement the phase correction at the higher frequency consists of the following steps:
[0128] 1) A regular grid (16) is defined on the reflecting surface, and on it an area is selected where the phase correction will be carried out, which could be the entire surface or the circular or elliptical crown (17) located around the edge of the antenna.
[0129] 2) An algorithm is defined to generate a random distribution of cells in which a 180o phase change will be made at the higher frequency, without altering the phase at the lower frequency. A possible way to implement this algorithm would be to perform the phase inversion in the cells of the grid that meet the following condition:
[0133] where d is the distance from the point in the center of the graticule to the edge of the reflector, C the width of the crown where the phase inversion cells are placed, r an adjustment coefficient (real and positive), the exponent to a coefficient that is a real number, positive or negative, and that is used to modify the variation of cell density with phase change, and rand (0,1) a random number between 0 and 1. As cases Note that if a = 1, the density of points grows linearly as we approach the edge; if a = 0, the point density is uniform throughout the crown; if a> 1, the density of points decreases more rapidly towards the center; if a <0, the point density grows from the edge towards the center of the antenna. Finally, note that if C is the distance from the center of the antenna to a point on the edge, the distribution of cells with phase correction would extend to the entire antenna.
[0134] ) Once the phase inversion is introduced in the cells that satisfy the above condition, the radiation patterns in the two frequency bands are calculated and compared with the patterns before correction and with the predefined specifications for the radiation patterns in each frequency.
[0135] ) If the diagram specifications are not fulfilled in both frequencies, the algorithm parameters are modified, and steps 2) and 3) are repeated, until reaching some diagrams that meet the required characteristics in both frequency bands, such as , have the same beamwidth and the same gain.
[0136] ) Next, the geometry of the conductive elements of each cell is defined and their dimensions are adjusted in such a way as to generate a phase change of 180 ° in the upper frequency band, without altering the phases in the lower band. For the design of the inverting conductive elements, initially a fixed angle of incidence is chosen, which is the one that corresponds to the center of the antenna.
[0137] ) Next, the conductive elements whose dimensions have been adjusted in step 5) are arranged in the cells defined according to the algorithm described in step 2) and the radiation patterns in both frequency bands are calculated. Having used the same dimensions of the conductive elements, which have been designed for a fixed angle of incidence, in all cells that invert the phase, a certain error will be obtained that will depend on the position of the cell in the antenna, since the angle of incidence changes according to its position on the grid. Therefore, the radiation pattern, in general, will not fully comply with the previously defined specifications, and will present certain discrepancies with respect to the patterns obtained in point 4.
[0138] 7) To correct these errors, the dimensions of all the conductive elements used to perform the phase inversion are optimized, using an optimization routine, such as the one based on the Davidon-Fletcher-Powell method, which calls iteratively to a routine electromagnetic modeling of the cell in periodic environment. The routine for optimizing the dimensions of the conductive elements aims to achieve a phase shift of 180 ° in the upper frequency band, without altering the phase in the lower band.
[0139] 8) Once the optimization process is finished, all the dimensions of all the printed conductive elements will be available, including those that invert the phase and those that do not invert it, and a CAD file will be generated for their manufacture by conventional photoengraving techniques. .
[0141] In some cases, it may be difficult to introduce a 180 ° phase shift at the higher frequency without modifying the phase at the lower frequency. In this case, two types of cells will be defined, which we will call cell type 1 and cell type 2, so that the phases of cells type 1 (18; 25; 31) at the lower and upper frequencies are those that correspond to the design of the antenna without phase correction (for a reflector they will be very similar phase values); while type 2 cells (19; 26; 32) will present a phase difference with respect to type 1 cells, which will be zero in the lower frequency and 180 ° in the upper frequency band (phase inversion). In this case, the entire lattice on the surface of the reflector will contain conductive elements, corresponding to cells of one type or another. These cells will be used in steps 6) and 7).
[0142] The same design procedure can be extended when the antenna has to work in three or more frequency bands. In this case, all design stages will be repeated for each frequency band, starting with the lowest and ending with the highest. That is, after having designed the antenna for the two lower frequencies, according to the technique described above, all the steps would be repeated for the next higher frequency, selecting and optimizing elements that introduce a 180 ° phase shift in said higher frequency and that alter the phase as little as possible at lower frequencies. Operation in more than two frequency bands requires more complex cells with a greater number of conductive elements and optimization processes that impose conditions in more frequency bands, implying a greater consumption of computational resources.
[0144] Phase correction at higher frequencies can be applied both in conventional reflectors (Fig. 1), as well as in “reflectarray” antennas on flat surfaces (Fig. 2), parabolic (Fig. 3), cylindrical, spherical or any type of surface. conformed. In the following, three possible preferred embodiments are described in detail.
[0146] A first embodiment of the invention consists of a parabolic reflector, with a diameter of approximately 2 meters, for a communications satellite that generates a multi-beam coverage in the band called Ka, which transmits in the band 17.5 to 20, 2 GHz and receives in the 29 to 30 GHz band, with equal beamwidth values (0.65 °) for the same gain level (46 dBi) in the upstream (29-30 GHz) and downstream (17 , 5-20.2 GHz), obtained thanks to a phase correction for the higher frequencies carried out at the edge of the antenna. Figure 1 shows a schematic drawing of this embodiment. The reflector is manufactured with the conventional technology used for antennas with space applications, which consists of using a mold with the parabolic surface, typically manufactured in invar to maintain dimensional stability in the curing processes, on which skins of 50 micron thick Kapton, coated with copper, which contains engraved conductive elements without phase inversion (14) in type 1 cells (18) and phase inverting conductive elements (15) in type 2 cells (19) in a crown of the outer perimeter. Several layers of a dielectric material are glued to the Kapton, which can be Kevlar skins, or quartz, impregnated in resin. On top of the dielectric layers, several layers of resin-impregnated carbon fiber will be glued, which will act as a conductor, and then there will be a core in the shape of a honeycomb that can be made of Kevlar, with the required thickness. to give sufficient rigidity to the structure. Finally, other structural layers will be additionally glued, according to the thermal and mechanical design of the structure. Although the surface of the reflector could be made of carbon fiber, which is conductive, in this case we have chosen to cover the surface with a Kapton skin with copper, so that the Kapton remains on the outside as a radome, to protect the copper layer. In the central part of the reflector, the copper will be continuous, or with a pattern of conductive elements distributed in a regular lattice, so that they introduce the same phase over the entire surface for the two frequency bands, while in an outer corona ( 17) the phase inverter conductive elements (15) will be inserted, arranged according to a random distribution in some cells of the lattice and that introduce the phase change from 180o to 29.5 GHz with respect to the phase introduced by the conductive elements in the rest of the grid.
[0148] Figure 4 shows the different layers of the reflector for a set of 3x3 cells with period Px = 4 mm, Py = 4 mm that includes cells of type 1 (18) and type 2 (19), whose conductive elements are square patches ( 14, 15) with sides li = 0.5 mm and l2 = 2.4 mm, respectively, recorded on a layer of flexible material (43), which for subsequent simulations has been chosen as Kapton 50 microns thick. The Kapton sticks to a layer of dielectric material (44), which can be made up of several fabrics of quartz fiber with epoxy resin with a thickness that in this case has been taken equal to h = 0.5 mm. Finally, the dielectric layer is glued to a layer of conductive material (45), which can be formed by several carbon fiber fabrics, that act as a conductive material. The figure also includes a core (46) and an additional layer of carbon fibers (47), which are intended to give the antenna structural rigidity.
[0150] Figure 5 shows the phase curves of the reflection coefficient, defined as the quotient between the reflected field and the incident field, for a field incident at an angle of 20o. It can be seen that cell type 1 introduces a phase of 151o at 19.7 GHz and a phase of 136o at 29.5 GHz, while for cell type 2, with larger conductive patches, which are used to perform the Phase inversion at 29.5 GHz, the phase is from 140o to 19.7 GHz and from -51o to 29.5 GHz. It can be seen that the phase difference between both types of cells is only 11o to 19.7 GHz, and is from 187o to 29.5 GHz, values close to 0o and 180o, respectively, with errors less than 11o, which are admissible, since they will not have a significant effect on the radiation patterns. It must be taken into account that typical errors around 10o are obtained due to manufacturing tolerances.
[0152] Figure 6 shows the radiation patterns in the XZ plane, defined according to Figure 1, corresponding to a conventional reflector made of carbon fiber with an aperture diameter of 1.8 m, at the frequencies of 19.7 GHz and 29.5 GHz. The reflector is fed by a horn antenna of circular section that has an aperture diameter of 55 mm, whose radiation pattern has symmetry of revolution and is modeled as a function cosq (0), with q = 24 to 19.7 GHz and q = 42 to 29.5 GHz, where 0 is the angle with respect to the axis of the horn. The diagrams are represented in gain (dBi), referred to the gain of an isotropic antenna, and include the illumination, ohmic reflector, and spillover losses (known as “spillover” in English). The figure also includes in dashed lines the specification masks that must be met, which consist of a minimum gain of 46 dBi in a 0.65o beamwidth, and maximum levels of 26 dBi for the side lobes in the zones coincident with the contiguous beams that have the same polarization and frequency, in order to ensure a Carrier / Interference ratio (called C / l by the acronym in English) of 20 dB. The gain is 3 dB higher in the frequency of the upstream beam (53.2 dBi at 29.5 GHz) than that of the downward beam (50.2 dBi at 19.7 GHz). From these diagrams it is also obtained that the beamwidth at a level 4 dB below the maximum is 0.65 ° at 19.7 GHz and 0.5 ° at 29.5 GHz. This inequality between the down beams and ascending will be corrected by the technique proposed in this invention.
[0154] Figure 7A shows the distribution of phases introduced by the reflector at the frequency of 29.5 GHz, when a phase correction is introduced in 50% of the cells of a regular grating of 5 mm side randomly distributed in a corona of width C = 30 cm measured from the edge of the reflector. Figure 7B shows the XZ plane gain radiation patterns obtained for the above phase distributions. It can be seen that the same gain value is achieved at both frequencies, but a 2 dB elevation is obtained in the level of the first secondary lobes at 29.5 GHz with respect to the lobes at the transmission frequency (19.7 GHz) , and an increase of 10 dB with respect to the first secondary lobe of the antenna without phase correction at 29.5 GHz (Fig. 6).
[0156] Figure 8A shows the distribution of phases introduced by the reflector at the frequency of 29.5 GHz, and Figure 8B, the radiation patterns in gain in the XZ plane, when phase reversing conductive elements are introduced that change the phase 180 ° in the cells of an outer crown of width 54 cm randomly distributed, with increasing density in the radial direction from the inner zone of the crown to the edge of the reflector. 180 ° phase shift at 29.5 GHz has been introduced in í d V cells
[0157] ^{of the lattice that satisfy the condition:} rl j ^{rand (®d) > 0.5 ^ s¡enc | 0 r =}
[0158] 0.6, C = 54 cm and a = 1.1. In Figure 8B it can be seen that the main lobe is practically identical at the frequencies of 19.7 and 29.5 GHz. Furthermore, at 29.5 GHz a reduction in the level of secondary lobes of 6.5 dB is obtained with respect to to the diagrams at 19.7 GHz, with which, this implementation of the invention substantially improves the level of side lobes obtained to 29.5 GHz in the implementation of Figure 7B. Furthermore, in the new implementation, the carrier / interference (C / l) ratio in reception improves to 27.4 dB, which is a value much higher than the specifications in communications satellites, and which cannot be achieved with other available techniques. in the prior state of the art. This very significant reduction in the level of secondary lobes has been possible thanks to the gradual increase in cell density with conductive elements that introduce phase inversion.
[0160] Figure 9A shows the distribution of phases introduced by the reflector at 29.5 GHz, and Figure 9B, the radiation patterns in gain in the XZ plane, when phase inverting conductive elements are introduced over the entire surface of the reflector, with an increasing density from the center to the edge of the reflector, defined by the parameters r = 3 and a = 3. Also in this case similar values of gain and beamwidth are achieved at the frequencies of 19.7 and 29.5 GHz, together with an even greater reduction in the level of secondary lobes, these being below 7 dBi, that is, 10 dB below the levels at the frequency of 29.5 GHz for the antenna in which the antenna has not been performed. phase correction (Fig. 6), the diagram of which has also been included in Fig. 9B.
[0162] The previously presented phase reversal cell distributions are just one example, but many more can be used with different parameter values or other parameterization criteria, allowing great flexibility to improve different aspects of the radiation pattern to the higher frequency. As an example, if the phase is reversed in all the cells of one half of the antenna, a difference pattern would be achieved in a principal plane, with a null in the direction of the paraboloid axis at the receiving frequency. If a 180o phase change is introduced in all the cells of two facing quadrants, a difference diagram in the two main planes would be obtained, which could be used to identify the direction of arrival of the wave.
[0163] In a second embodiment, a flat "reflectarray" type antenna is considered, as shown in Figure 2, with dimensions of approximately 1.8 meters in diameter. The antenna is designed for a communications satellite that generates multi-beam coverage in the so-called Ka band. The antenna transmits in the 19.2 to 20.2 GHz band and receives in the 29 to 30 GHz band, with equal beamwidth values (0.65 °) for the same gain level (46 dBi) in the upward (29-30 GHz) and downward (19.2-20.2 GHz) beams, obtained thanks to a phase correction for the upper frequency band carried out in a corona on the edge of the “reflectarray”. In this case, the "reflectarray" is designed according to the prior state of the art to generate a collimated beam in the two frequency bands, using a plurality of phase shifting cells (25), but without phase inversion, with printed conductive elements. (21, 22), arranged in a regular grid on the surface of the "reflectarray". As commented, the regular grid (16) has been highlighted with a broken line in a small elliptical area, but it extends to the entire surface of the “reflectarray”, as well as the phase shifting cells (25) that were represented in an elliptical area. (20), but that extend to the entire surface of the “reflectarray”. In this example, the phase shifting cells (25) of the "reflectarray" are made up of two layers of dielectric material (60, 67), with conductive elements made up of printed conductive dipoles (21, 22) in each layer to control the phase of the field. reflected independently in each frequency band and in each polarization. The dipoles (21) are oriented along the X axis, and are used to adjust the component of the reflected electric field along X, while the dipoles (22) are oriented along the Y axis, and are used to adjust the component of the electric field reflected according to Y. The dimensions of the dipoles are adjusted in all the cells of the “reflectarray”, considering only the type of phase-shifting cells without phase inversion (25) or cells type 1, using optimization routines together with electromagnetic simulation routines to generate the collimated beams in both frequency bands and for both linear polarizations. The "reflectarray" can operate in double polarization, both linear and circular, which will be the same polarization in which the feeders used operate.
[0164] To achieve equal gain and beamwidth in the two frequency bands, some of the cells of the “reflectarray” designed to focus the beam, the so-called type 1 cells (25), will be replaced by type 2 cells (26) that introduce a 180o phase shift in the upper frequency band (29-30 GHz), without altering the phases in the lower band. Type 2 cells (26) are formed by phase inverting conductive elements (23, 24), which invert the phase at the higher frequency, and will be located according to a random distribution in a corona near the edge of the "reflectarray". Note that in this case, in Figures 2 and 10, the conductive elements (23, 24) that invert the phase in the cells (26) are smaller than the conductive elements (21, 22) that do not invert the phase in the cells (26). cells (25), but it does not necessarily have to be the case, being the characteristic that differentiates them, that their dimensions have been adjusted to introduce a phase difference of 180o in the field reflected at the higher frequency, with respect to the phase introduced by elements (21, 22) without phase inversion.
[0166] Figure 10 shows a perspective view of a set of "reflectarray" cells that can be used in this embodiment. The “reflectarray” cells generally consist of two orthogonal sets of five parallel dipoles (50, 51, 52, 53, 54; 55, 56, 57, 58, 59) printed on a lower dielectric layer, called A (60), and two additional sets of three parallel dipoles (61, 62, 63; 64, 65, 66) stacked on top of the first sets and printed on top of a second upper dielectric sheet designated B (67). The period, in this case, has been chosen as Px = Py = 6.5 mm, which is equal to 0.66 times the wavelength at the highest frequency (29.5 GHz). This period ensures that diffraction lobes do not appear for angles of incidence less than 30o. In each set of 3 or 5 dipoles, the lateral dipoles located on both sides of the central one are of the same length to maintain low levels of cross-polar radiation. Note that the lengths of the dipoles in the direction of the X axis, Ai, Ia ² , Ia ³ , Ibi, Ib ² , are adjusted to control the phases of the electric field incident with polarization X, while the lengths of the printed dipoles in the direction of the Y axis, Ia ⁴ , Ias, Ia6, Ib ³ , Ib ⁴ , are adjusted to control the phases of the electric field that it affects Y polarization. This type of cell allows independent phase control in the transmission (Tx) and reception (Rx) frequencies, based on the lengths of the dipoles: the dipoles of the upper layer, called B (67), they will not disturb the phase response at 19.7 GHz, since they are shorter than those of the lower layer, while the dipoles of the lower layer A (60), will behave like a ground plane at 29.5 GHz for the dipoles of the upper layer B. Therefore, the lengths of the dipoles can be adjusted first in the lower layer (60) to produce the phase required at 19.7 GHz to focus the beam, and then, those of the upper layer (67) to provide the required phase at the upper frequency of 29.5 GHz, including the phase shift at the upper frequency in type 2 cells (26). The advantage of this type of “reflectarray” cell is that it allows implementing different phases for each frequency and for each polarization.
[0168] A periodic “array” of cells such as those shown in figure 10 is considered, with period Px = Py = 6.5 mm, two layers of dielectric (Diclad 880) with dielectric constant zrA = £ Ve = 2.17 and tangent of losses tan <5¿ = tan <5e = 0.001, and thicknesses for each layer: hA = 1.5 mm and hB = 1 mm. The sets of dipoles oriented in the X and Y directions are identical, the width of the dipoles being equal to w = 0.25 mm, the separations between the centers of the adjacent parallel dipoles in each layer: S ^xa = S ^ya = 0, 5 mm and S ^xb = S ^{and b} = 1 mm, and the relationships between the lengths of the central and lateral dipoles of each cluster: / ^a 1 = 0.65- / ^a 3, / ^a 2 = 0.8- / ^a 3, / ^a 4 = 0.65- / ^a 6, / ^a 5 = 0.8- / ^a 6, where / ^a 3 and / a6 are the lengths of the central dipoles in the lower layer (60), and / b1 = 0.8 / B2, / B3 = 0.8 / B4, where / B2 and / B4 are the lengths of the central dipoles in the upper layer (67). The phase values of the reflection coefficient are calculated at the frequencies of Tx (19.7 GHz) and Rx (29.5 GHz) when the dimensions of the dipoles of both layers ( / A ³ , / B2) are varied in such a way independent for X polarization, and the results are depicted in Figures 11A and 11B , respectively. These figures show that the phase at the 19.7 GHz frequency is proportional to the lengths of the dipoles of layer A (/ A3), while the phase at the higher frequency (29.5 GHz) depends mainly on the length. / ^b 2 of the upper layer (67), as mentioned previously. These results show that the lengths of the dipoles in both layers can be adjusted to achieve any phase value independently at each frequency, in a range of 360 °. In this way, it is possible to implement the phase correction technique described in this invention, since for predetermined phase values at each frequency (19.7 and 29.5 GHz), the lengths of the dipoles that generate said predetermined phase values (cells type 1, 25), and also the dimensions of another cell (type 2, 26) that generates the same phase at the lower frequency (19.7 GHz) and a phase shift of 180 ° can be obtained at the higher frequency (29.5 GHz) with respect to that of the type 1 cell, by adjusting the dimensions of the dipoles in the upper layer (67). In Figures 11A and 11B, the points corresponding to an increasing phase identical to the two frequencies (type 1 phase shift cells) have been marked with a white line with black circles, and the lines marked with squares and diamonds have also been included, which correspond to type 2 cells, which maintain the same phase at the lower frequency (19.7 GHz) and introduce a phase shift of -180 ° (squares) or 180 ° (diamonds) at the upper frequency (29.5 GHz) with with respect to the phase of type 1 cells (marked with circles). Assuming that the same phase distribution is to be implemented in both frequencies, first the “reflectarray” would be designed using type 1 cells (center line marked with circles), and later, to perform the phase correction at the higher frequency, the cells Randomly selected according to the algorithm described above, they will be replaced by other cells in any of the lines marked with squares or diamonds (type 2 cells), which produce a phase inversion at the higher frequency. The dimensions of the dipoles are equal to or less than half a wavelength in the higher frequency band, and equal to or less than 0.3 wavelengths in the lower frequency band.
[0170] Unlike the first embodiment, in this case the dimensions of the phase inverting conductive elements must be calculated independently for each position of the cell in the "reflectarray", since they have to introduce a phase shift of 180 ° at the frequency higher than what the designed reflectarray would have without phase correction.
[0172] Figures 12A and 12B show the phase distributions that the reflectarray must introduce to collimate the beam at the frequencies of 19.7 GHz and 29.5 GHz, respectively. Note that the phase distribution at 29.5 GHz has a greater number of phase jumps than 360 °, because the diameter of the “reflectarray” is 177 wavelengths at 29.5 GHz, and 118 wavelengths at 19 , 7 GHz. Figure 12C shows the radiation patterns in the XZ plane (defined as shown in Figure 2), corresponding to a 1.8 m diameter flat "reflectarray". The "reflectarray" has been designed to produce a collimated beam at the frequencies of 19.7 GHz and 29.5 GHz, using the cells (25) described above (Fig. 10). It is fed by a horn antenna (12) of circular section that has an aperture diameter of 55 mm, whose radiation pattern has symmetry of revolution and is modeled as a cosq (0) function, with q = 24 to 19.7 GHz, and q = 42 to 29.5 GHz. The gain at 29.5 GHz is 53.2 dBi, while at 19.7 GHz it is 50.4 dBi, that is, 2.8 dB higher in the frequency of the upward beam relative to that of the downward beam. The beamwidth at a level 4 dB below the maximum is 0.65 ° at 19.7 GHz and 0.5 ° at 29.5 GHz. This mismatch between the down and up beams will be corrected by the technique proposed in this invention.
[0174] Figure 13A shows the distribution of phases introduced by the "reflectarray" at the frequency of 29.5 GHz when the phase is reversed at 29.5 GHz in cells randomly distributed in an outer crown of width 54 cm, with increasing density in the radial direction from the inner zone of the crown towards the edge of the reflector. 29.5 GHz phase reversed on
[0176] cells of the grid whose center meets the condition: r 1 | rand { 0,1)> 0.5
[0177] where r = 0.6, C = 54 cm and a = 1.1. Figure 13B shows the radiation patterns in the XZ plane, which would be obtained for a "reflectarray" with the phase distribution shown in Figure 12A at 19.7 GHz and that shown in Figure 13A at 29.5 GHz. Similar to the first embodiment based on a conventional reflector, it can be seen that, with the phase correction proposed in this invention, the main lobe is practically identical at the frequencies 19.7 and 29 , 5 GHz, maintaining a low level of secondary lobes, with a C / l ratio better than 26 dB at the frequency of 29.5 GHz.
[0179] A third preferred embodiment (figure 3) consists of a parabolic “reflectarray” type antenna, which operates in double circular polarization (to the right and to the left), transmitting and receiving both circular polarizations in different bands with the appropriate phase correction to obtain the same. beamwidth and gain values in both frequency bands. The parabolic reflectarray is powered by a circular horn that transmits and receives the two circular polarizations in the two bands. The parabolic surface is defined according to the prior state of the art to focus the beam in the lower frequency band. , in the same way as in the first embodiment. In figure 3, an elliptical broken line (30) has been represented, in which a plurality of type 1 (31) and type 2 (32) phase shifting cells are included, but as in the case of reference (20) in the second embodiment, it represents the distribution of the phase shifting cells (31) and (32) over the entire surface of the reflectarray. In this case the plurality of cells (30) comprises two types of cells (31, 32) of the “reflectarray” that are found in a regular grid (16) on the surface of the “reflectarray”, where the phase-shifting cells of type 1 (31), which appear in the figure with dipoles of larger dimensions are used for c Forming the beam without phase inversion, and type 2 phase shifting cells (32), with smaller dipoles, are used to invert the phase at the higher frequency. The phase shifting cells of both types (31, 32) contain two groups of printed conductive elements, in this case a first group made up of two symmetrical arcs (34A, 34B) and a second group made up of sets of dipoles (33, 35), each of them having a predetermined angle of rotation, with which the phase of the reflected field in circular polarization is adjusted independently for each frequency band, by means of the known technique as sequential rotation. According to this technique, when a right-hand circularly polarized field (RHCP) hits the cells of the reflectarray, propagating in the direction of the negative Z axis, given by the expression:
[0182] Eme = E0 ( x 'jy') t (2)
[0183] / V | / V |
[0184] where E0 is the amplitude of the incident field, x> y, the unit vectors along the axes of the element rotated an angle amt. The electric field reflected on the surface of the “reflectarray”, which propagates in the direction of the positive Z axis, will be:
[0186] ¿W = 2 e'2 "" £ s («xl. - R ^ x '- jy ^ U ^ Rrr) ( x' + jy ') _ (3) where Rxx and Ryy are the reflection coefficients of the components of field according to the x 'and y' directions of the rotated elements. When the condition Rxx ^{- - R yy} is fulfilled, the reflected field is circularly polarized to the right and with a phase proportional to twice the angle of rotation of the conducting element, as is shown in the following expression:
[0188] ~ RHCP
[0189] Wef - eJ2 «rr, _R xx _{E 0 (X} - _{jy 1)} (4)
[0191] However, when the incident field is polarized to the left and the condition Rxx ^{- - R yy} is met, the reflected field will be circularly polarized to the left and will undergo a phase change of equal magnitude to that of equation (4) and opposite sign . In the case under consideration, in which the antenna operates in two frequency bands, the phase shifting cells (31, 32) are formed by two groups of conductive elements, a first group of two arcs (34A, 34B) printed on the layer B (67) and a second group of dipoles (33) containing 3 dipoles in layer A (60) and another 3 dipoles perpendicular to the first ones printed in layer B (67), whose rotation angles determine the phase of the field reflected in each frequency band. For the property Rxx ^{- - R yy} to be fulfilled, it is necessary to design the printed conductive elements in such a way that the two linear components of the field reflected in the coordinate system of the rotated element (electric field in the x 'and y' directions) differ by 180 °, for each frequency band. Taking into account that the introduced phase is of opposite sign in each circular polarization, in this embodiment the cells of the "reflectarray" are rotated sequentially to deflect the reflected beams in opposite directions in each circular polarization. As an example, consider an antenna with a diameter of 1.8 meters that generates beams with a beam width at -4 dB of 0.65 ° in the transmission band from the satellite (17.5-20.2 GHz). In transmission, the beams are deflected 0.28 ° for right polarization and -0.28 ° for left polarization, by rotating the conductive elements formed by two symmetrical arcs (34A, 34B), generating a single antenna two proximal beams 0.56 ° apart in orthogonal circular polarization, for each feeder operating in double circular polarization. For this application, the groups of dipoles without phase inversion (33) that control the phase in the receive band (29-30 GHz) are rotated in the opposite direction, to produce a deviation of -0.28 ° for the polarization to right and 0.28 ° for left polarization, receiving in each feeder of the same antenna, the signals of two adjacent coverage areas that operate in orthogonal circular polarization. Note that the transmitting and receiving beams for each coverage area operate in orthogonal polarizations, to minimize interference. Taking into account that each feeder generates two beams, the number of antennas on the satellite, which are currently four reflectors, can be cut in half. Said antenna is designed by techniques known in the prior state of the art, but it would have the problem of presenting different beamwidths in reception and transmission. To solve this drawback, some of the conductive elements of the second group, formed by a set of dipoles (33) of the "reflectarray", designed to generate the adjacent beams in the left and right polarizations, will be replaced in the cells selected according to a random distribution by phase inverting conductive elements (35), formed by dipoles of different dimensions, which introduce a 180 ° phase change in the upper frequency band (29-30 GHz), without altering the phases in the lower band. The conductive elements (35) that invert the phase at the higher frequency, they have been represented in FIG. 14C with dipoles in the Y direction (72 ', 73', 74 ') of smaller dimensions (lY1, lY2). The phase inverting conductive elements (35) at the higher frequency will be located according to a random distribution in a corona near the edge of the “reflectarray”, which is defined by the parameters r, d, C and a of equation (1) shown previously.
[0193] Figures 14A and 14B show a perspective view and a top view of the "reflectarray" cell used in this embodiment, consisting of two stacked dielectric layers, a lower layer (60) and an upper layer (67), and two levels of metallization. To operate in two different frequency bands by sequential rotation technique, the cell contains a group of conductive elements printed for each frequency. The first group of conductive elements is formed by two symmetrical arcs (34A, 34B) with a rotation angle aArc with which the phase is controlled at the lower frequency (19.7 GHz); while the second group of conductive elements is formed by three parallel dipoles (72, 73, 74) printed on layer B (67) and three other dipoles perpendicular to the first ones (75, 76, 77) printed on layer A ( 60), with a rotation angle aD¡P that allows to adjust the phase at the higher frequency (29.5 GHz).
[0195] The selection of the materials, the period and the thickness of the two dielectric layers are set by the prior state of the art, depending on the operating frequencies. For the case considered here, two layers of dielectric (Diclad 880B) of thickness 1.524 mm (layer A, 60) and 0.127 mm (layer B, 67) have been used with a dielectric constant er = 2.17, and a tangent of losses rang = 0.001, the period being 6.5 mm. The arcs have a width of 0.2 mm and an internal radius of 2.65 mm, and their length is adjusted so that in all cells of the “reflectarray”, with the corresponding angle of rotation, a difference of phase between the components of the reflected field in the rotated system, formed by the axes ^{(x 'Arc,} and ^Are), of 180 °. On the other hand, in each group of 3 parallel dipoles (72, 73, 74; 75, 76, 77) in each layer the lengths of the lateral dipoles located remain the same to the sides of the central dipole, and these lengths are scaled relative to that of the central dipole. In this case the scale factor in each layer is 0.77, that is, I ^x 2 = 0.77- lXi, Iy ² = 0.77- lYi. The dipoles are 0.4 mm wide and have a 1.2 mm separation between the centers of adjacent parallel dipoles. Once these parameters are set, the lengths of the dipoles in each layer are independently adjusted to control the phase that is introduced at the higher frequency (29.5 GHz) in each component of the reflected field along the axes of the rotated dipole system. (x'Dip, y'Dip). In this case, any phase value can be achieved, as long as a 180o difference is maintained between the phases of both field components. This allows a further adjustment in the phase distribution of the reflected field at the higher frequency, allowing the radiation pattern to be modified at that frequency, for example, to reduce the level of secondary lobes, shape or change the direction of the beam, without modifying the diagrams at the lower frequency.
[0197] Taking into account that in this embodiment the phase in each frequency band is adjusted by means of the independent rotation of group 1 of arcs (34A, 34B) for the lower frequency, and of group 2 of dipoles (72, 73, 74; 75, 76, 77) for the upper one, and that we also have the adjustment of the lengths of the dipoles to independently control the phase of each linear component of the electric field in the directions of the rotated dipoles, the phase inversion at the upper frequency in some conductive elements of the second group (35) it can be done by changing 180o the phase of the reflection coefficient of the linear field components in the
[0198] rotated system
[0199] element, as can be seen in the expression:
[0200] + RHCP
[0201] E'tf ( f2) = e, 2 "* - RX. _{Dip ^ dip} E, ( x '- jy ^{' DJ} (4)
[0203] Z2 being the higher frequency. It must be taken into account that, for the previous expression to be fulfilled, that is, for the orthogonal polarization component to be canceled in expression (2), the following must be fulfilled condition: ^ xDipxDip ^ yd¡pyd¡p . For this condition to be fulfilled there are two
[0204] options, with different sign (180 ° phase shift) in the coefficient of the x'Dip component in the system of rotated dipoles. That is, the condition that
[0205] ^{/ DI _ / D} 180 ^°
[0206] fulfill both options is: XdpXdp ^ ^ X d ^ Xd ^ - , the cells being
[0208] with reflection coefficient dxUpXdíp those associated with type 1 (31), cells without
[0210] phase correction; while cells with coefficient ^{D I} xDpxDp are
[0211] associate type 2, (32), cells with phase inversion. This last type of cell has been represented in figure 14C, in which the dimensions I'yi, I'y ² of the dipoles (72 ', 73', 74 ') printed on layer B (67) and the dimensions I'xi, I'x ² of the dipoles (75 ', 76', 77 ') printed on the A layer (60) have been modified to achieve phase inversion in the upper frequency band. Figures 15A and 15B show the phases of the components according to x'Dip and y'Dip of the
[0212] reflection coefficients ( RxD¡pxD¡p 'R ^yd¡py d¡¡1 ) respectively at 29.5 GHz when the dimensions of the dipoles of both layers ( Ixi, Iyi) are varied in the form D
[0213] Independent. These graphs show that the phase of DxD¡pxD¡p at the frequency
[0214] 29.5 GHz fundamentally depends on the length IXi of the dipoles in the x'D¡p direction . In the two graphs, the points at which increasing the lengths of both sets of dipoles is obtained a phase variation of the coefficient D have been represented with a central line marked with circles
[0215] dxd¡pxd¡p greater than 360 ° and the phase difference between both is maintained
[0216] 180 ° coefficients ( RxD¡pxD¡p _ ~ R ^yd ¡ ^{p and} d¡¡, ). In addition to this line, another line has been marked with squares, whose points continue to fulfill the same previous condition, and also present a change in the sign of the coefficient in the
[0217] direction x'Dip, R'xD¡pxD¡p = ~ RxD¡px D¡p, that is, it changes the phase 180 ° with respect to the values of the line marked with circles. In this way, to invert the phase in the components at the higher frequency, it is enough to go from a point from the center line to another on the line marked with squares, where R
[0218] we have a 180 ° phase shift in the xapxap coefficient. Note that the
[0219] points of both lines (the one marked with circles and the one marked with squares)
[0220] they fulfill the condition in RxDpxDp ^{= ~ R ydpy} d¡¡ ^,, and in addition there is a change of sign in the X and Y components of the reflection coefficients R'xDpxDp = ~ RxDpxDp and R ^dpyd p = ~ R ^ydpyd p , when moving from one line to another. Therefore, either of the two lines can be chosen to select cells (31) without phase inversion, so that cells with phase inversion (32) would be on the other line. By way of example, in the cell that reverses phase (32) at the higher frequency represented in figure 14C, the second group of conductive elements (35) has the dipoles (75 ', 76', 77 ') in the layer A of greater lengths ( l'xi, I'x ² ) and the dipoles (72 ', 73', 74 ') in layer B of shorter lengths ( I'yi, I'y ² ), with respect to the elements in phase shifting cell (31) without phase inversion, represented in figure 14B. That is, it has been considered that the cells (31) that do not invert the phase are those corresponding to the line marked with squares in Figures 15A-B, while the cells (32) that invert the phase correspond to the lines marked with circles.
[0222] Figure 16A shows the right-hand circular polarization phase distribution, at the frequency of 29.5 GHz, introduced by a 1.8 m diameter “reflectarray” designed to generate two 0.65 ° wide beams. 0.56 ° spaced beams in orthogonal circular polarizations, when illuminated with a horn operating in double circular polarization in the Tx (19.7 GHz) and Rx (29.5 GHz) frequency bands, using the cells described above (Fig. 14A-C). The “reflectarray” is fed by a horn antenna with a circular section having an aperture diameter of 55 mm, whose radiation pattern has symmetry of revolution and is modeled as a cosq (0) function, with q = 24 to 19, 7 GHz and q = 42 to 29.5 GHz. To achieve the same beamwidth in the two frequency bands, a D
[0223] 180 ° phase shift at 29.5 GHz in the xapxap coefficient in some cells
[0224] randomly distributed in an outer crown 54 cm wide, with increasing density in the radial direction from the inner zone of the crown to the edge of the reflector. The distribution of cells where the change is introduced
[0225] ( d V
[0226] 180 ° phase is given by the condition J> 0.5. s¡enc | 0 ^
[0227] the distance from the point on the graticule to the edge of the reflector, C the width of the crown where the phase correction is introduced and a = 1.2. The phase distribution introduced for the orthogonal polarization (circular to the left) will have the opposite sign to that shown in Figure 16A, in order to deflect the beam in the opposite direction.
[0229] Figure 16B shows the radiation patterns in the XZ plane (defined in Figure 3) at both frequencies (19.7 and 29.5 GHz) and for right-hand (RHCP) and left-hand (LHCP) circular polarizations. Note that the beams at the frequencies of Tx and Rx are deflected 0.28 ° in opposite directions for each circular polarization. Additionally, the beam in each direction at the Tx and Rx frequencies has orthogonal polarizations, to minimize interference. In the figure it can be seen that both the gain and the beamwidth are approximately equal in both frequencies and polarizations, which shows that the phase correction technique proposed in this invention has worked satisfactorily also in this third embodiment. It can be seen that the gain and beamwidth are very close to those obtained in the case of a conventional reflector with phase correction (first embodiment), which means that the losses introduced by the "reflectarray" are very small. The values obtained by simulation give us losses of 0.3 dB in the Tx band and 0.4 dB in the Rx band, which are of the same order as those that would be obtained in a carbon fiber reflector.
[0231] The design techniques used in the three described embodiments have a number of steps in common, as described above. Figure 17 shows a flow chart detailing all the steps to follow to design an antenna with multi-frequency phase correction. In a first step, the specifications (79) of the radiation patterns in the operating frequency bands are defined, based on the selected application. Second, the phase-corrected antenna (80) is designed by techniques known in the prior state of the art, to operate in previously defined frequency bands. The antenna designed without phase correction, as seen in the previous examples, presents different radiation patterns at each frequency. Assuming that the antenna operates in only two frequency bands, for simplicity, the different characteristics of the radiation patterns at both frequencies, such as directivity, level of secondary lobes and beamwidth, can be simultaneously improved at both frequencies following the technique of phase correction proposed in this invention, following the following steps:
[0232] a) Definition of a regular grid on the reflecting surface (81). b) The positions and density of the cells in the regular grid (82) in which the phase will be inverted in the upper frequency band are obtained, according to a random distribution obtained by means of an algorithm, such as the one described above, but which it could be another. c) The phase is inverted in the upper frequency band in the selected cells and the radiation patterns at both frequencies are calculated (83).
[0233] d) The patterns obtained before and after the phase correction performed in step c) are compared with the predefined specifications for the radiation patterns in each frequency band (84).
[0234] e) If the diagram specifications are not met in both frequency bands, the parameters of the algorithm used to adjust the distribution of cells in which the phase is inverted are modified, and steps b), c) and d) are repeated. ; repeating the process until specifications are met in both frequency bands.
[0235] It must be taken into account that up to now the ideal phases have been used on the surface of the antenna, on which the phase in the upper frequency band, without performing any electromagnetic simulation, which makes this part of the process extremely fast. It is from now on when it is necessary to define the conductive elements capable of inverting the phase in the upper frequency band, without modifying the phase of the reflected field in the lower band. Results have been presented previously for three different types of antenna, which do not have to be the only ones. The phase inverting conductive elements will have a different geometry, depending on the type of antenna in which we want to carry out the phase correction. For example, in the case of a reflector, it is enough to use small square patches, while in the parabolic reflectarray of the third embodiment the elements used to invert the phase are much more complex.
[0236] f) Once the cells that must invert the phase in the upper frequency have been defined, for the type of antenna selected, the geometry will be defined and the dimensions of the conductive elements capable of inverting the phase in the upper frequency band will be adjusted, without alter the phases in the lower frequency bands (85). At this point, a fixed angle of incidence is considered, the one that corresponds to the center of the reflecting surface, so that it is only necessary to design a cell with phase inversion.
[0237] If the antenna must operate in an additional upper frequency band (86A), the new upper and lower frequencies (86B) are redefined and the above process must be repeated independently for each upper frequency band, continuing with the following steps:
[0238] g) The new positions and cell density are obtained in the lattice to invert the phase of the reflected field for the new upper frequency band (82), without affecting the phase distribution in the lower bands.
[0239] h) The radiation patterns in all the frequency bands considered are calculated after introducing the phase inversions in the selected cells (83), and compared with the diagrams before correction and with the predefined specifications for radiation patterns in each frequency band (84).
[0240] If the diagram specifications are not met in all the considered frequency bands (84),
[0241] i) the parameters used to adjust the distribution of cells in which the phase inversion is performed for the new upper frequency band (82) are modified and steps g) (82) and h) are repeated (83-84 ).
[0242] j) The geometry is defined and the dimensions of the phase inverting conductive elements are adjusted in the new upper frequency band, without altering the phases in the lower frequency bands, considering a fixed angle of incidence (85).
[0243] Once the previous steps have been carried out for all frequency bands (86A-B), we continue with the steps that involve optimization and electromagnetic simulation. This is a process that requires more computational cost, but it is only done on a reduced number of elements, which is where we invert the phase in the higher frequency bands. This process is carried out through the following steps: k) The conductive elements that carry out the phase inversion in each frequency band are arranged and the radiation patterns are calculated in all the frequency bands (87), carrying out an electromagnetic simulation of all the elements, considering that they are in a periodic environment. Since initially a phase inverting conductive element has been defined for each higher frequency, without taking into account that the angle of incidence varies with the position of the element on the reflecting surface, it is usual that the diagram specifications are not fully met radiation in all frequency bands, therefore it is necessary to carry out the next step.
[0244] l) Optimize the dimensions of all the conductive elements used to perform the phase inversion (88), by means of an optimization routine that iteratively calls a routine of electromagnetic modeling of the elements in a periodic environment, until the specifications are met in all frequency bands (89).
[0245] m) From the dimensions of all the conductive elements obtained in the optimization process, the masks (90) are generated for the photoengraving manufacturing of the dielectric layers with the optimized conductive elements.
[0247] The first and third embodiments make it possible to reuse the same parabolic surface mold in the manufacture for satellite antennas of different missions, considerably reducing the recursive costs of the antenna system. In addition to the three detailed preferred embodiments, the proposed phase correction can also be used to maximize gain at several frequencies simultaneously, in high-gain antennas for data transmission in space probes used in scientific missions, such as the Mars Express missions. , Venus Express, Bepi Colombo, which use different frequency bands.
[0249] INDUSTRIAL APPLICATION
[0250] The fundamental application of this invention is in antennas embedded in satellites, both for communications and scientific missions, which have to transmit and receive signals in different frequency bands. It is a reflector antenna for sending and receiving data between the satellite and ground stations. The antenna is built with materials and manufacturing technologies qualified for space, such as pre-impregnated carbon fiber fabrics with resins (CFRP in the English literature), to which is added one or more pre-impregnated layers of dielectric fibers, such as Kevlar or quartz, and Kapton with gravure printed metallizations. This technology has been developed and used in the dichroic sub-reflectors of various scientific missions, such as Cassini, Voyager, Mars-Express, Venus-Express, Bepi Ccolombo, etc.
[0251] An application of interest is the optimization of the gain simultaneously at several frequencies for high gain antennas that are used to send data to the earth from space, as for example, in the S (2 2.4 GHz), X bands (7-8 GHz), Ka (32-34 GHz). This technique can compensate for differences in the position of the feeder phase center at each frequency.
[0253] Another application of interest is in multi-beam antennas to generate cellular coverage for access to broadband satellite “internet”. The present invention makes it possible to obtain the same performance for the ascending and descending beams, without the need to shape the surface of the antenna. For this application, both conventional reflectors, to which layers are added in the manufacturing process, and "reflectarray" type antennas can be used. In the case of “reflectarray” antennas, which can be flat or concave, in addition to ensuring the same behavior of the antenna in the two frequency bands, it is possible to reduce the number of antennas on the satellite by half, using two “reflectarrays "Instead of four reflectors, since each" reflectarray "antenna is capable of generating two contiguous beams in orthogonal polarization for each feeder.
[0255] Finally, the independent phase correction technique at various frequencies allows the same antenna to be used for several missions, such as, for example, to generate shaped beams at a certain frequency and multiple beams in another band. Therefore, this type of phase-corrected reflector has an important field of application in the space industry as an alternative to the different types of reflectors embedded in satellites, parabolic, grating or shaped beam.

权利要求:
Claims (14)
[1]
1. Reflecting antenna that transmits and receives signals in at least two frequency bands, which are called by their ordinal, so that the first band corresponds to the lower frequency, the second band to the immediately higher frequency, and so on , with multi-frequency phase correction to improve radiation patterns, comprising a reflector (10) and a primary feeder (12) configured to illuminate the reflector, the reflector being formed by a multilayer structure (11) formed by one or more layers of conductive and dielectric materials (44; 60, 67), including conductive elements (14, 15; 21, 22, 23, 24; 33, 34A-B, 35) distributed on the surface of at least one layer of dielectric material, (44; 60, 67), characterized by comprising at least a part of the printed conductive elements (15; 23, 24; 35) on the surface of at least one layer of dielectric material (44; 60, 67) , configured to invert the phase of the reflex field Adopted in at least one of the higher order frequency bands in which the antenna operates, without producing an effect on the phase of the reflected field in the first band; said conducting elements, called phase inverting conducting elements, comprising dimensions equal to or less than half a wavelength in said at least one higher-order frequency band in which the phase is inverted, and at the same time equal to or less than 0 , 3 wavelengths in the immediately lower frequency band, the position and distribution of said phase inverting conductive elements being defined by a random distribution with a variable density that adjusts to form a radiation pattern in at least one frequency band order without modifying the radiation pattern in the first frequency band, to improve the radiation patterns in all frequency bands.
[2]
2. A reflector antenna with multi-frequency phase correction according to claim 1, characterized by comprising the elements conductors (21, 22, 23, 24; 33, 34A-B, 35) distributed in cells (18, 19; 25, 26; 31, 32) in a regular lattice on the surface of at least one layer of dielectric material ( 60, 67), where said conductive elements comprise a shape, size and angle of rotation with respect to the grating previously calculated to generate an initial distribution of phases in the reflected field necessary to collimate or shape the radiated beam in all transmission bands and antenna reception; and where a part of the cells of the lattice, comprises phase inverting conductive elements (23, 24; 35), configured to carry out a 180o phase change in the higher order frequency bands, with respect to the initial phases generated to collimate or shape the beam.
[3]
3. Reflecting antenna for circular polarization with multifrequency phase correction according to claim 2, characterized by including a first group of conductive elements (34A-B) and a second group of conductive elements (33, 35) in each cell (31; 32 ) of a regular lattice on the surface of at least one layer of dielectric material (60, 67), in which a rotation angle of the first group of conductive elements (34A-B) is used to adjust the phase in the first band frequency and an angle of rotation of the second group (33, 35) of conductive elements to adjust the phase in the second frequency band, using the technique known as sequential rotation independently for each group of conductive elements, comprising the conductive elements of each group a shape and size previously calculated to introduce a 180o phase change between the two components of the electric field reflected by the conductive element in each of the operating frequency bands of the antenna, the conductive elements of each group being rotated in the lattice at angles equal to half the phase required in each operating frequency band of the antenna to collimate or shape the beam in circular polarization when a feeder is used that operates in circular polarization in several bands of frequency, where a part of the cells (32) of the lattice, comprise the second group of conductive elements (35), which invert the phase of the reflected field in circular polarization in the second frequency band, with respect to the one initially calculated for collimate or shape the beam.
[4]
4. A reflector antenna for circular polarization with multifrequency phase correction according to claim 3, characterized in that the first group of conductive elements comprises two symmetrical arcs (34A, 34B) with a rotation angle aArc used to control the phase in the first frequency band, and the second group of conductive elements (33) comprises two orthogonal sets of three parallel dipoles (72, 73, 74; 75, 76, 77), each of them printed on a dielectric layer, with a rotation angle Odip, which is used to adjust the phase in the second frequency band.
[5]
5. Multi-frequency phase corrected reflector antenna according to any of claims 1 to 4, characterized in that it comprises the conductive elements that invert the phase in the frequency bands in which the antenna operates, excluding the first band, distributed only in outer crowns, defining a central zone free of phase correction, and each outer crown and each configuration of phase inverting conductive elements being defined for each frequency band in which the correction is made.
[6]
6. A reflector antenna with multi-frequency phase correction according to claim 5, characterized in that it comprises the conductive elements that invert the phase in the frequency bands in which the antenna operates, excluding the first band, distributed in random positions within a or several outer crowns, the random distribution of the elements with phase correction in each crown being defined by an algorithm that includes at least two adjustment parameters, independently for each frequency band in which the correction is made.
[7]
7. A reflector antenna with multi-frequency phase correction according to any of claims 1 to 4, characterized by having the conductive elements that invert the phase to the frequency bands in which the antenna operates, excluding the first band, distributed in a manner random with increasing density from the center of the antenna to the edge.
[8]
8. A reflector antenna with multi-frequency phase correction according to any of claims 1 to 7, characterized in that it has a reflective surface selected from a flat, parabolic, spherical or cylindrical surface.
[9]
9. Multi-frequency phase-corrected reflector antenna according to any of claims 1 to 8, characterized by comprising qualified materials for space applications, selected from carbon fiber pre-impregnated with resins, Kapton coated with copper, Kapton-Germanium coated with copper, quartz fibers pre-impregnated with low loss resins and qualified adhesives for space and combination applications.
[10]
10. A reflector antenna with multi-frequency phase correction according to claim 8, characterized by having a parabolic reflecting surface, configured to generate a multi-cellular coverage from a geostationary satellite with a phase correction in the second frequency band, which is used for reception, calculated to obtain the same beamwidth and the same gain in the first frequency band, which is used for transmission, and in the second frequency band, used for reception.
[11]
11. Design method to obtain a reflective antenna that transmits and receives signals in at least two frequency bands, which are called by their ordinal, so that the first band corresponds to the lower frequencies, the second band to the immediately higher frequencies, and so on, including the antenna conducting elements (14, 15; 21, 22, 23, 24; 33, 34A, 34B, 35) distributed in cells (18 , 19; 25, 26; 31, 32) on the surface of at least one layer of dielectric material, (44; 60, 67), said method comprising a first step, defining specifications (79) of the radiation patterns in the antenna operating frequency bands, ordered from the lower frequency (first band) to the higher frequency band (fn), a second step of the antenna design without phase correction (80) that generates the diagrams radiation in the directions defined according to the preset specifications (79) in all frequency bands; characterized in that the method also comprises:
a) define a regular grating on the reflecting surface of the antenna and define the first frequency band as the lower band (//, / = 1), (81);
b) obtain some positions and density of cells in the regular grid (82) in which the phase will be inverted in the immediately superior frequency band (// + 1), according to a selection of random distribution that is a function of the distance from the cell to the edge of the reflector and which is adjusted with at least two parameters;
c) in the cells selected in the previous point, the phase for the reflected field in the immediately higher frequency band is inverted, without changing the phase in any of the lower frequency working bands, and the radiation patterns are calculated in the two frequency bands, called lower (/ ■) and immediately higher (// + 1), after introducing the phase inversion in the selected cells (83);
d) compare the radiation patterns obtained after the phase correction performed in step c) with the patterns obtained in the second step before inverting the phase in the selected cells, and with the predefined specifications in the first step for the diagrams radiation at each frequency (84); if the diagram specifications are not met in both frequency bands;
e) modifying the parameters used to adjust the distribution of cells in which the phase is inverted, obtaining new cell positions with phase inversion at the immediately higher frequency (82) and repeating steps c) and d);
f) define the geometry and adjust the dimensions of the conductive elements in each cell previously selected to invert the phase in the immediately higher frequency band (/ ¡+ 1), without altering the phase in the lower frequency band ( f¡) , for an angle of incidence that corresponds to the center of the reflecting surface (85);
if the antenna must operate in an additional higher frequency band (86A), the previous immediately higher band is redefined as the lower band, and the new band is considered as the immediately higher frequency (86B);
g) obtaining new positions and density of cells in the lattice according to a random distribution with a density that depends on the distance of the cell from the edge of the reflector and that is adjusted with at least two parameters; to introduce in said cells an inversion of the phase of the reflected field for the new immediately higher frequency band (82), without affecting the phase distribution in the lower bands;
h) calculate the radiation patterns in all considered frequency bands after introducing the phase inversions in the selected cells (83), compare them with the patterns before phase correction and with the predefined specifications for the radiation patterns in each frequency band (84);
if the diagram specifications are not met in all considered frequency bands;
i) modify the parameters used to adjust the distribution of cells in which the phase inversion is performed for the band of immediately higher frequencies (82) and repeat steps g) (82) and h) (83-84);
j) define the geometry and adjust the dimensions of the phase inverting conductive elements in the immediately higher frequency band, without altering the phases in the lower frequency bands, for the angle of incidence that corresponds to the center of the antenna (85) ;
once the previous steps have been carried out for all frequency bands (86A-B), it comprises;
k) arranging the conductive elements defined above in the positions obtained to perform the phase inversion in each frequency band, and calculate the radiation patterns in all the frequency bands (87);
l) optimize the dimensions of all the conductive elements used to perform the phase inversion (88), through a conventional optimization routine that iteratively calls an electromagnetic modeling routine of the conductive elements in a periodic environment, until the specifications are met in all frequency bands (89);
m) from the dimensions of all the conductive elements obtained in the optimization process, masks (90) are generated for the photoengraving of the dielectric layers with the optimized conductive elements
[12]
Design method according to claim 11, to obtain an antenna according to claims 1 and 2, characterized in that it comprises as a step prior to step b): adjusting the dimensions of the conductive elements distributed in a regular grid on the surface of the al minus one layer of dielectric material, to generate the phase distribution in the reflected field necessary to collimate or shape the radiated beam in all the transmission and reception bands of the antenna; adjusting in steps from j) to), only the dimensions of a part of the conductive elements of the grid to introduce the Phase inversion in higher order frequency bands, with respect to the phase initially calculated in the second step to collimate or shape the beam.
[13]
Design method according to claim 11 to obtain an antenna according to claims 3 or 4 that operates in circular polarization, characterized in that it comprises as a step prior to step b): defining the geometry and adjusting the dimensions of at least a first group of conductive elements (34A-B) and a second group (33, 35) in each cell of a uniform lattice on the surface of at least one layer of dielectric material, to introduce a 180o phase change between the two components of the field electric reflected by the conductive element in each of the antenna operating bands; adjust the angle of rotation of the first and second group of conductive elements (34A-B, 33, 35) using the technique known as sequential rotation to obtain the angles of rotation of the elements of the first group (34A-B) to shape the beam to the lower frequency band, and the rotation angles of the elements of the second group (33, 35) and of the following groups, if any, to form the beam in the second frequency band, and in the following bands, respectively, when the feeder operates in circular polarization at various frequencies;
adjust in steps from j) to), the dimensions of the conducting elements of the second group (35) and of the following groups, if any, used to control the phase in the second frequency band and subsequent ones, if any, to introduce a phase inversion in the reflected field in circular polarization in the second frequency band and subsequent ones, if any, with respect to the phase initially calculated in the second step to collimate or shape the beam in each frequency band.
[14]
Design method according to any one of claims 11 to 13, characterized in that the cells selected to introduce a change 180 ° phase lines for higher-order frequency bands are found only in an outer crown, which is defined for each frequency band.

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公开号 | 公开日

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ES2791798B2|2021-03-08|

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ES202030819A|ES2791798B2|2020-07-31|2020-07-31|REFLECTOR ANTENNA WITH MULTI-FREQUENCY PHASE CORRECTION FOR SPACE APPLICATIONS AND DESIGN METHOD OF THE SAME.|ES202030819A| ES2791798B2|2020-07-31|2020-07-31|REFLECTOR ANTENNA WITH MULTI-FREQUENCY PHASE CORRECTION FOR SPACE APPLICATIONS AND DESIGN METHOD OF THE SAME.|